Circuit for testing the max capabilities of tl494. Charger for car battery on TL494

Automotive Charger or an adjustable laboratory power supply with an output voltage of 4 - 25 V and a current of up to 12A can be made from an unnecessary computer AT or ATX power supply.

Let's look at several scheme options below:

Options

From a computer power supply with a power of 200W, you can actually get 10 - 12A.

AT power supply circuit for TL494

Several ATX power supply circuits for TL494

Rework

The main modification is as follows: we unsolder all the extra wires coming from the power supply to the connectors, leave only 4 pieces of yellow +12V and 4 pieces of black housing, twist them into bundles. We find on the board a microcircuit with number 494, in front of the number there may be different letters DBL 494, TL 494, as well as analogues MB3759, KA7500 and others with a similar connection circuit. We are looking for a resistor going from the 1st leg of this microcircuit to +5 V (this is where the red wire harness was) and remove it.

For an regulated (4V - 25V) power supply, R1 should be 1k. Also, for the power supply, it is desirable to increase the capacity of the electrolyte at the 12V output (for a charger it is better to exclude this electrolyte), make several turns on a ferrite ring with a yellow beam (+12V) (2000NM, 25 mm in diameter is not critical).

It should also be borne in mind that on the 12 volt rectifier there is a diode assembly (or 2 back-to-back diodes) rated for a current of up to 3 A, it should be replaced with the one on the 5 volt rectifier, it is rated up to 10 A, 40 V , it is better to install the BYV42E-200 diode assembly (Schottky diode assembly Ipr = 30 A, V = 200 V), or 2 back-to-back powerful diodes KD2999 or similar ones in the table below.

If you need to connect the soft-on pin to the common wire to start the ATX power supply (the green wire goes to the connector). The fan needs to be turned 180 degrees so that it blows inside the unit, if you are using it as a power supply, it is better to power the fan with the 12th the legs of the microcircuit through a 100 Ohm resistor.

It is advisable to make the case from dielectric, not forgetting about the ventilation holes; there should be enough of them. Original metal case, use at your own risk.

It happens that when you turn on the power supply at a high current, the protection may work, although for me it doesn’t work at 9A, if anyone encounters this, you should delay the load when turning it on for a couple of seconds.

Another interesting option for redesigning a computer power supply.

In this circuit, voltage (from 1 to 30 V) and current (from 0.1 to 10A) are adjusted.

Voltage and current indicators are well suited for a homemade unit. You can buy them on the Trowel website.

P O P U L A R N O E:

When I go out by car, I take my laptop with me...

One day I came across an article on an amateur radio website about how to make a car adapter for a laptop.

A simple circuit (see below) - one microcircuit and a pair of transistors...

SWITCH POWER SUPPLY FOR TL494 AND IR2110

Most automotive and network voltage converters are based on a specialized TL494 controller, and since it is the main one, it would be unfair not to briefly talk about the principle of its operation.
The TL494 controller is a plastic DIP16 package (there are also options in a planar package, but it is not used in these designs). Functional diagram controller is shown in Fig. 1.

Picture 1 - Structural scheme TL494 chips.

As can be seen from the figure, the TL494 microcircuit has very developed control circuits, which makes it possible to build converters on its basis to suit almost any requirements, but first a few words about the functional units of the controller.
ION circuits and protection against undervoltage. The circuit turns on when the power reaches the threshold of 5.5..7.0 V (typical value 6.4V). Until this moment, the internal control buses prohibit the operation of the generator and the logical part of the circuit. Current idle move at supply voltage +15V (output transistors are disabled) no more than 10 mA. ION +5V (+4.75..+5.25 V, output stabilization no worse than +/- 25mV) provides a flowing current of up to 10 mA. The ION can only be boosted using an NPN emitter follower (see TI pp. 19-20), but the voltage at the output of such a “stabilizer” will greatly depend on the load current.
Generator generates a sawtooth voltage of 0..+3.0V (the amplitude is set by the ION) on the timing capacitor Ct (pin 5) for the TL494 Texas Instruments and 0...+2.8V for the TL494 Motorola (what can we expect from others?), respectively, for TI F =1.0/(RtCt), for Motorola F=1.1/(RtCt).
Allowable operating frequencies from 1 to 300 kHz, with the recommended range Rt = 1...500 kOhm, Ct = 470pF...10 μF. In this case, the typical temperature drift of frequency is (of course, without taking into account the drift of attached components) +/-3%, and the frequency drift depending on the supply voltage is within 0.1% over the entire permissible range.
For remote shutdown generator, you can use an external key to short-circuit the Rt input (6) to the ION output, or short-circuit Ct to ground. Of course, the leakage resistance of the open switch must be taken into account when selecting Rt, Ct.
Rest phase control input (duty factor) through the rest phase comparator sets the required minimum pause between pulses in the arms of the circuit. This is necessary both to prevent through current in the power stages outside the IC, and for stable operation of the trigger - the switching time of the digital part of the TL494 is 200 ns. The output signal is enabled when the saw exceeds the voltage at control input 4 (DT) by Ct. At clock frequencies up to 150 kHz with zero control voltage, the resting phase = 3% of the period (equivalent bias of the control signal 100..120 mV), at high frequencies the built-in correction expands the resting phase to 200..300 ns.
Using the DT input circuit, it is possible to set a fixed resting phase ( R-R divider), soft start mode (R-C), remote shutdown (key), and also use DT as a linear control input.
The input circuit is assembled using PNP transistors, so the input current (up to 1.0 μA) flows out of the IC rather than into it. The current is quite large, so high-resistance resistors (no more than 100 kOhm) should be avoided. See TI, page 23 for an example of surge protection using a TL430 (431) 3-lead zener diode. Error Amplifiers
When using an RC frequency-dependent OS, you should remember that the output of the amplifiers is actually single-ended (series diode!), so it will charge the capacitance (upward) and will take a long time to discharge downward. The voltage at this output is within 0..+3.5V (slightly more than the generator swing), then the voltage coefficient drops sharply and at approximately 4.5V at the output the amplifiers are saturated. Likewise, low-resistance resistors in the amplifier output circuit (feedback loop) should be avoided.
Amplifiers are not designed to operate within one clock cycle of the operating frequency. With a signal propagation delay inside the amplifier of 400 ns, they are too slow for this, and the trigger control logic does not allow it (side pulses would appear at the output). IN real circuits The PN cutoff frequency of the OS circuit is selected on the order of 200-10000 Hz.
Trigger and output control logic - With a supply voltage of at least 7V, if the saw voltage at the generator is greater than at the DT control input, and if the saw voltage is greater than at any of the error amplifiers (taking into account the built-in thresholds and offsets) - the circuit output is allowed. When the generator is reset from maximum to zero, the outputs are switched off. A trigger with paraphase output divides the frequency in half. With logical 0 at input 13 (output mode), the trigger phases are combined by OR and supplied simultaneously to both outputs; with logical 1, they are supplied in phase to each output separately.
Output transistors - npn Darlingtons with built-in thermal protection (but without current protection). Thus, the minimum voltage drop between the collector (usually closed to the positive bus) and the emitter (at the load) is 1.5 V (typical at 200 mA), and in a circuit with a common emitter it is a little better, 1.1 V typical. The maximum output current (with one open transistor) is limited to 500 mA, the maximum power for the entire chip is 1 W.
Switching power supplies are gradually replacing their traditional relatives in audio engineering, since they look noticeably more attractive both economically and in size.
The same factor that switching power supplies contribute significantly to the distortion of the amplifier, namely the appearance of additional overtones, is no longer relevant mainly for two reasons - the modern element base makes it possible to design converters with a conversion frequency significantly higher than 40 kHz, therefore the power modulation introduced by the power supply will already be in ultrasound. In addition, a higher power supply frequency is much easier to filter, and the use of two L-shaped LC filters along the power supply circuits already sufficiently smoothes out the ripples at these frequencies.
Of course, there is a fly in the ointment in this barrel of honey - the difference in price between a typical power supply for a power amplifier and a pulsed one becomes more noticeable as the power of this unit increases, i.e. The more powerful the power supply, the more profitable it is in relation to its standard counterpart.
And that is not all. When using switching power supplies, it is necessary to adhere to the rules for installing high-frequency devices, namely the use of additional screens, feeding the power part of the common wire to the heat sinks, as well as correct ground wiring and connection of shielding braids and conductors.

After a short lyrical digression about the features of switching power supplies for power amplifiers, the actual circuit diagram of a 400W power supply: Picture 1. Schematic diagram
switching power supply for power amplifiers up to 400 W

ENLARGE IN GOOD QUALITY
After the TL494 controller there is an IR2110 half-bridge driver, which actually controls the gates of the power transistors.
The use of the driver made it possible to abandon the matching transformer, which is widely used in computer power supplies. The IR2110 driver is loaded onto the gates through the R24-VD4 and R25-VD5 chains that accelerate the closing of the field gates. Power keys
VT2 and VT3 operate on the primary winding of the power transformer. The midpoint required to obtain alternating voltage in the primary winding of the transformer is formed by elements R30-C26 and R31-C27.
A few words about the operating algorithm of the switching power supply on the TL494:
At the moment of supplying a mains voltage of 220 V, the capacitances of the primary power supply filters C15 and C16 are infected through resistors R8 and R11, which does not allow the diol bridge VD to be overloaded by a short circuit current of completely discharged C15 and C16.
At the same time, capacitors C1, C3, C6, C19 are charged through a line of resistors R16, R18, R20 and R22, stabilizer 7815 and resistor R21.

As soon as the voltage on capacitor C6 reaches 12 V, the zener diode VD1 “breaks through” and current begins to flow through it, charging capacitor C18, and as soon as the positive terminal of this capacitor reaches a value sufficient to open thyristor VS2, it will open.

It should be noted here that the duration of the soft start is limited, since the current passing through resistors R16, R18, R20, R22 is not enough to power the TL494 controller, the IR2110 driver and the switched-on relay winding - the supply voltage of these microcircuits will begin to decrease and will soon decrease to a value at which TL494 will stop generating control pulses. And it is up to this moment that the soft start mode must be completed and the converter must return to normal operating mode, since the TL494 controller and IR2110 driver receive their main power from a power transformer (VD9, VD10 - midpoint rectifier, R23-C1-C3 - RC filter, IC3 - 15 V stabilizer) and that is why capacitors C1, C3, C6, C19 have such large ratings - they must maintain the controller’s power supply until it returns to normal operation.
The TL494 stabilizes the output voltage by changing the duration of control pulses of power transistors at a constant frequency - Pulse-Width Modulation - PWM. This is only possible if the value of the secondary voltage of the power transformer is higher than that required at the output of the stabilizer by at least 30%, but not more than 60%.

Figure 3. Operating principle of a PWM stabilizer.

As the load increases, the output voltage begins to decrease, the optocoupler LED IC1 begins to glow less, the optocoupler transistor closes, reducing the voltage on the error amplifier and thereby increasing the duration of the control pulses until the effective voltage reaches the stabilization value (Figure 3).
It should be noted that the TL494 controller does not regulate the duration of each pulse depending on the output voltage, but only the average value, i.e. the measuring part has some inertia. However, even with capacitors installed in the secondary power supply with a capacity of 2200 μF, power failures at peak short-term loads do not exceed 5%, which is quite acceptable for HI-FI class equipment. We usually install capacitors in the secondary power supply of 4700 uF, which gives a confident margin for peak values, and the use of a group stabilization choke allows us to control all 4 output power voltages.
The pulse block The power supply is equipped with overload protection, the measuring element of which is the current transformer TV1. As soon as the current reaches a critical value, thyristor VS1 opens and bypasses the power supply to the final stage of the controller.
The control pulses disappear and the power supply goes into standby mode, which it can remain in for quite a long time, since the thyristor VS2 continues to remain open - the current flowing through resistors R16, R18, R20 and R22 is enough to keep it in the open state. How to calculate a current transformer.
To exit the power supply from standby mode, you must press the SA3 button, which will bypass the thyristor VS2 with its contacts, the current will stop flowing through it and it will close.
As soon as the contacts SA3 open, the transistor VT1 closes itself, removing power from the controller and driver. Thus, the control circuit will switch to minimum consumption mode - thyristor VS2 is closed, therefore relay K1 is turned off, transistor VT1 is closed, therefore the controller and driver are de-energized.
When assembling more powerful options, you should pay attention to the capacitors of the primary power supply smoothing filters C15 and C16. The total capacitance of these capacitors must be proportional to the power of the power supply and correspond to the proportion 1 W of the output power of the voltage converter corresponds to 1 µF of the capacitance of the primary power filter capacitor. In other words, if the power of the power supply is 400 W, then 2 capacitors of 220 μF should be used, if the power is 1000 W, then 2 capacitors of 470 μF or two of 680 μF must be installed.
This requirement has two purposes. Firstly, the ripple of the primary supply voltage is reduced, which makes it easier to stabilize the output voltage. Secondly, using two capacitors instead of one facilitates the operation of the capacitor itself, since electrolytic capacitors The TK series are much easier to obtain, and they are not entirely intended for use in high-frequency power supplies - the internal resistance is too high and these capacitors will heat up at high frequencies. Using two pieces, the internal resistance is reduced, and the resulting heating is divided between two capacitors.
When used as power transistors IRF740, IRF840, STP10NK60 and similar ones (for more information about the transistors most commonly used in network converters, see the table at the bottom of the page), diodes VD4 and VD5 can be abandoned altogether, and the values ​​of resistors R24 and R25 can be reduced to 22 Ohms - power The IR2110 driver is quite enough to control these transistors. If a more powerful switching power supply is being assembled, then more powerful transistors. Attention should be paid to both the maximum current of the transistor and its dissipation power - switching stabilized power supplies are very sensitive to the correct installation of the snubber and without it, the power transistors heat up more because currents formed due to self-induction begin to flow through the diodes installed in the transistors. Read more about choosing a snubber.
Also, the closing time that increases without a snubber makes a significant contribution to heating - the transistor stays in linear mode longer.
Quite often they forget about one more feature of field-effect transistors - with increasing temperature, their maximum current decreases, and quite strongly. Based on this, when choosing power transistors for switching power supplies, you should have at least a two-fold maximum current reserve for power amplifier power supplies and a three-fold reserve for devices operating on a large, unchanging load, for example, an induction smelter or decorative lighting, powering low-voltage power tools.
The output voltage is stabilized using the group stabilization choke L1 (GLS). You should pay attention to the direction of the windings of this inductor. The number of turns must be proportional to the output voltages. Of course, there are formulas for calculating this winding unit, but experience has shown that the overall power of the core for a DGS should be 20-25% of the overall power of the power transformer. You can wind until the window is filled by about 2/3, not forgetting that if the output voltages are different, then the winding with a higher voltage should be proportionally larger, for example, you need two bipolar voltages, one at ±35 V, and the second to power the subwoofer with voltage ±50 V.
We wind the DGS into four wires at once until 2/3 of the window is filled, counting the turns. The diameter is calculated based on a current intensity of 3-4 A/mm2. Let's say we got 22 turns, let's make up the proportion:
22 turns / 35 V = X turns / 50 V.
X turns = 22 × 50 / 35 = 31.4 ≈ 31 turns
Next, I’ll cut two wires for ±35 V and wind up another 9 turns for a voltage of ±50.
ATTENTION! Remember that the quality of stabilization directly depends on how quickly the voltage changes to which the optocoupler diode is connected. To improve the stabilization coefficient, it makes sense to connect an additional load to each voltage in the form of 2 W resistors with a resistance of 3.3 kOhm. The load resistor connected to the voltage controlled by the optocoupler should be 1.7...2.2 times less.

The circuit data for network switching power supplies on ferrite rings with a permeability of 2000 Nm are summarized in Table 1.

However, it is not always possible to recognize the brand of ferrite, especially if it is ferrite from horizontal transformers of televisions. You can get out of the situation by finding out the number of turns experimentally. More details about this in the video:

Using the above circuitry of a switching power supply, several submodifications were developed and tested, designed to solve a particular problem at various powers. The printed circuit board drawings for these power supplies are shown below.
Printed circuit board for a switching stabilized power supply with power up to 1200...1500 W. Board size 269x130 mm. In fact, this is a more improved version of the previous one. printed circuit board. It is distinguished by the presence of a group stabilization choke, which allows you to control the magnitude of all power voltages, as well as an additional LC filter. Has fan control and overload protection. The output voltages consist of two bipolar power sources and one bipolar low-current source, designed to power the preliminary stages.

Appearance PCB power supply up to 1500 W. DOWNLOAD IN LAY FORMAT

A stabilized switching network power supply with a power of up to 1500...1800 W can be made on a printed circuit board measuring 272x100 mm. The power supply is designed for a power transformer made on K45 rings and located horizontally.

It has two bipolar power sources, which can be combined into one source to power an amplifier with two-level power supply and one bipolar low-current source for preliminary stages.

Printed circuit board of a switching power supply up to 1800 W. DOWNLOAD IN LAY FORMAT This power supply can be used to power automotive equipment. high power

External view of the printed circuit board of the power supply for automotive equipment DOWNLOAD IN LAY FORMAT

The power supply up to 2000 W is made on two boards measuring 275x99, located one above the other. The voltage is controlled by one voltage. Has overload protection.

The file contains several options for the “second floor” for two bipolar voltages, for two unipolar voltages, for the voltages required for two and three level voltages.

The power transformer is located horizontally and is made on K45 rings.

Appearance of a “two-story” power supply DOWNLOAD IN LAY FORMAT

A power supply with two bipolar voltages or one for a two-level amplifier is made on a board measuring 277x154. Has a group stabilization choke and overload protection.

Appearance of a “two-story” power supply DOWNLOAD IN LAY FORMAT

The power transformer is on K45 rings and is located horizontally.

Appearance of a “two-story” power supply DOWNLOAD IN LAY FORMAT

Power up to 2000 W.

External view of the printed circuit board DOWNLOAD IN LAY FORMAT

Almost the same power supply as above, but has one bipolar output voltage.

The switching power supply has two power bipolar stabilized voltages and one bipolar low current.

Equipped with fan control and overload protection. It has a group stabilization choke and additional LC filters.

The switching power supply has two power bipolar stabilized voltages and one bipolar low current.

There is much more space for ferrites on boards than there could be. The fact is that it is not always necessary to go beyond the sound range. Therefore, additional areas are provided on the boards. Just in case, a small selection of reference data on power transistors and links to where I would buy them. By the way, I have ordered both TL494 and IR2110 more than once, and of course power transistors. It’s true that I didn’t take the entire assortment, but so far I haven’t come across any defects.

Who has not encountered in their practice the need to charge a battery and, disappointed in the lack of a charger with the necessary parameters, was forced to purchase a new charger in a store, or reassemble the necessary circuit?
So I have repeatedly had to solve the problem of charging various batteries, when there was no suitable memory at hand. Accounted for a quick fix collect something simple, in relation to a specific battery.

The situation was tolerable until the need for mass preparation and, accordingly, charging the batteries arose. It was necessary to produce several universal chargers - inexpensive, operating in a wide range of input and output voltages and charging currents.

The charger circuits proposed below were developed for charging lithium-ion batteries, but it is possible to charge other types of batteries and composite batteries (using the same type of cells, hereinafter referred to as AB).

All presented schemes have the following main parameters:
input voltage 15-24 V;
charge current (adjustable) up to 4 A;
output voltage (adjustable) 0.7 - 18 V (at Uin=19V).

All circuits were designed to work with power supplies from laptops or to work with other power supplies with DC output voltages from 15 to 24 Volts and were built on widespread components that are present on the boards of old computer power supplies, power supplies of other devices, laptops, etc.

Memory circuit No. 1 (TL494)

The inverse input (pin 2) of the same error amplifier is supplied with a comparison voltage, regulated by a variable resistor PR1, from a reference voltage source built into the chip (ION - pin 14), which changes the potential difference between the inputs of the error amplifier.
As soon as the voltage value on R10 exceeds the voltage value (set by the variable resistor PR1) at pin 2 of the TL494 microcircuit, the charging current pulse will be interrupted and resumed again only at the next cycle of the pulse sequence generated by the microcircuit generator.
By thus adjusting the width of the pulses on the gate of transistor VT2, we control the battery charging current.

Transistor VT1, connected in parallel with the gate of a powerful switch, provides the necessary discharge rate of the gate capacitance of the latter, preventing “smooth” locking of VT2. In this case, the amplitude of the output voltage in the absence of a battery (or other load) is almost equal to the input supply voltage.

With an active load, the output voltage will be determined by the current through the load (its resistance), which allows this circuit to be used as a current driver.

When charging the battery, the voltage at the switch output (and, therefore, at the battery itself) will tend to increase over time to a value determined by the input voltage (theoretically) and this, of course, cannot be allowed, knowing that the voltage of the battery being charged lithium battery should be limited to 4.1V (4.2V). Therefore, the memory uses a threshold device circuit, which is a Schmitt trigger (hereinafter - TS) on an op-amp KR140UD608 (IC1) or on any other op-amp.

When the required voltage value on the battery is reached, at which the potentials at the direct and inverse inputs (pins 3, 2 - respectively) of IC1 are equal, a high logical level (almost equal to the input voltage) will appear at the output of the op-amp, causing the LED indicating the end of charging HL2 and the LED to light up optocoupler VH1 which will open its own transistor, blocking the supply of pulses to output U1. The key on VT2 will close and the battery will stop charging.

Once the battery is charged, it will begin to discharge through the reverse diode built into VT2, which will be directly connected in relation to the battery and the discharge current will be approximately 15-25 mA, taking into account the discharge also through the elements of the TS circuit. If this circumstance seems critical to someone, a powerful diode (preferably with a low forward voltage drop) should be placed in the gap between the drain and the negative terminal of the battery.

The TS hysteresis in this version of the charger is chosen such that the charge will begin again when the voltage on the battery drops to 3.9 V.

This charger can also be used to charge series-connected lithium (and other) batteries. It is enough to calibrate the required response threshold using variable resistor PR3.
So, for example, a charger assembled according to scheme 1 operates with a three-section serial battery from a laptop, consisting of dual elements, which was mounted to replace the nickel-cadmium battery of a screwdriver.
The power supply from the laptop (19V/4.7A) is connected to the charger, assembled in the standard case of the screwdriver charger instead of the original circuit. The charging current of the “new” battery is 2 A. At the same time, transistor VT2, working without a radiator, heats up to a maximum temperature of 40-42 C.
The charger is switched off, naturally, when the battery voltage reaches 12.3V.

The TS hysteresis when the response threshold changes remains the same as a PERCENTAGE. That is, if at a shutdown voltage of 4.1 V, the charger was turned on again when the voltage dropped to 3.9 V, then in this case The charger is switched on again when the voltage on the battery drops to 11.7 V. But if necessary, the depth of hysteresis can be changed.

Charger Threshold and Hysteresis Calibration

Setting the current mode is even easier.
1. We turn off the threshold device using any available (but safe) methods: for example, by “connecting” the PR3 engine to the common wire of the device or by “shorting” the LED of the optocoupler.
2. Instead of the battery, we connect a load in the form of a 12-volt light bulb to the output of the charger (for example, I used a pair of 12V 20-watt lamps to set up).
3. We connect the ammeter to the break of any of the power wires at the input of the charger.
4. Set the PR1 engine to minimum (to the maximum left according to the diagram).
5. Turn on the memory. Smoothly rotate the PR1 adjustment knob in the direction of increasing current until the required value is obtained.
You can try to change the load resistance towards lower values ​​of its resistance by connecting in parallel, say, another similar lamp or even “short-circuiting” the output of the charger. The current should not change significantly.

During testing of the device, it turned out that frequencies in the range of 100-700 Hz were optimal for this circuit, provided that IRF3205, IRF3710 were used (minimum heating). Since the TL494 is underutilized in this circuit, the free error amplifier on the IC can be used to drive a temperature sensor, for example.

It should also be borne in mind that if the layout is incorrect, even a correctly assembled pulse device will not work correctly. Therefore, one should not neglect the experience of assembling power pulse devices, described repeatedly in the literature, namely: all “power” connections of the same name should be located at the shortest distance relative to each other (ideally at one point). So, for example, connection points such as the collector VT1, the terminals of resistors R6, R10 (connection points with the common wire of the circuit), terminal 7 of U1 - should be combined almost at one point or through a straight short and wide conductor (bus). The same applies to drain VT2, the output of which should be “hung” directly onto the “-” terminal of the battery. The terminals of IC1 must also be in close “electrical” proximity to the battery terminals.

Memory circuit No. 2 (TL494)

As you can see, in order to quickly change the current mode and work with different numbers of elements connected in series, fixed settings have been introduced with trimming resistors PR1-PR3 (setting the current), PR5-PR7 (setting the end of charging threshold for different quantities elements) and switches SA1 (selection of charging current) and SA2 (selection of the number of battery cells to be charged).
The switches have two directions, where their second sections switch the mode selection indication LEDs.

Another difference from the previous device is the use of a second error amplifier TL494 as a threshold element (connected according to the TS circuit) that determines the end of battery charging.

Well, and, of course, a p-conductivity transistor was used as a key, which simplified the full use of the TL494 without the use of additional components.

The method for setting the end of charging thresholds and current modes is the same, as for setting up the previous version of the memory. Of course, for a different number of elements, the response threshold will change multiples.

When testing this circuit, we noticed stronger heating of the switch on the VT2 transistor (when prototyping I use transistors without a heatsink). For this reason, you should use another transistor (which I simply did not have) of appropriate conductivity, but with better current parameters and lower resistance open channel, or double the number of transistors indicated in the circuit by connecting them in parallel with separate gate resistors.

The use of these transistors (in a “single” version) is not critical in most cases, but in this case, the placement of the device components is planned in a small-sized case using small radiators or no radiators at all.

Memory circuit No. 3 (TL494)

Charger circuit No. 3a (TL494)

Memory circuit No. 4 (TL494)

Such a module is always useful for bench tests of both batteries and other devices. It makes sense to use built-in devices (voltmeter, ammeter). Formulas for calculating storage and interference chokes are described in the literature. Let me just say that I used ready-made various chokes (with a range of specified inductances) during testing, experimenting with a PWM frequency from 20 to 90 kHz. I didn’t notice any particular difference in the operation of the regulator (in the range of output voltages 2-18 V and currents 0-4 A): minor changes in the heating of the key (without a radiator) suited me quite well. The efficiency, however, is higher when using smaller inductances.
The regulator worked best with two series-connected 22 µH chokes in square armored cores from converters integrated into motherboards laptops.

Memory circuit No. 5 (MC34063)

Memory circuit No. 6 (UC3843)

In schemes 5 and 6, the same sets of components (including chokes) were used in the experiments. According to the test results, all of the listed circuits are not much inferior to each other in the declared range of parameters (frequency/current/voltage). Therefore, a circuit with fewer components is preferable for repetition.

Memory circuit No. 7 (TL494)

A relay is used to switch the battery from the “charge” mode to the “load” mode.

Working with the memory is similar to working with previous devices. Calibration is carried out by switching the toggle switch to the “calibration” mode. In this case, the contact of the toggle switch S1 connects the threshold device and a voltmeter to the output of the integral regulator IC2. Having set the required voltage for the upcoming charging of a specific battery at the output of IC2, using PR3 (smoothly rotating) the HL2 LED lights up and, accordingly, relay K1 operates. By reducing the voltage at the output of IC2, HL2 is suppressed. In both cases, control is carried out by a built-in voltmeter. After setting the PU response parameters, the toggle switch is switched to charge mode.

Scheme No. 8

Charger design

Designed several, different in functionality and element base, digital pulse duration meters.

More than 30 improvement proposals for the modernization of units of various specialized equipment, incl. - power supply. For a long time now I have been increasingly involved in power automation and electronics.

Why am I here? Yes, because everyone here is the same as me. There is a lot of interest here for me, since I am not strong in audio technology, but I would like to have more experience in this area.

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CHARGING DEVICE FOR CAR BATTERIES

Another charger assembled according to the circuit of a key current stabilizer with a unit for monitoring the achieved voltage on the battery to ensure that it is turned off at the end of charging. A widely used specialized microcircuit is used to control the key transistor TL494 (KIA494, KA7500B, K1114UE4). The device provides charge current regulation within 1 ... 6 A (10A max) and output voltage 2 ... 20 V.

Key transistor VT1, diode VD5 and power diodes VD1 - VD4 through mica spacers must be installed on a common radiator with an area of ​​200 ... 400 cm2. Most important element in the circuit is a choke L1. The efficiency of the circuit depends on the quality of its manufacture. The requirements for its manufacture are described in You can use it as a core pulse transformer from the TV power supply 3USTST or similar. It is very important that the magnetic core has a slot gap of approximately 0.2 ... 1, 0 mm to prevent saturation at high currents. The number of turns depends on the specific magnetic circuit and can be in the range of 15 ... 100 turns of PEV-2 2.0 mm wire. If the number of turns is excessive, then a soft whistling sound will be heard when the circuit operates at rated load. As a rule, the whistling sound occurs only at medium currents, and with a heavy load, the inductance of the inductor due to the magnetization of the core drops and the whistling stops. If the whistling sound stops at low currents and with a further increase in the load current the output transistor begins to heat up sharply, then the area of ​​the magnetic circuit core is insufficient to operate at the selected generation frequency - it is necessary to increase the operating frequency of the microcircuit selection of resistor R4 or capacitor C3 or install a larger choke. With absence power transistor structures p-n-p in the circuit you can use powerful transistors of the structure n-p-n , as it shown on the picture.

The key transistor VT1, diode VD5 and power diodes VD1 - VD4 through mica spacers must be installed on a common radiator with an area of ​​200 ... 400 cm2. The most important element in the circuit is inductor L1. The efficiency of the circuit depends on the quality of its manufacture. As a core, you can use a pulse transformer from a 3USTST TV power supply or similar. It is very important that the magnetic core has a slot gap of approximately 0.5 ... 1.5 mm to prevent saturation at high currents. The number of turns depends on the specific magnetic circuit and can be in the range of 15 ... 100 turns of PEV-2 2.0 mm wire. If the number of turns is excessive, then a soft whistling sound will be heard when the circuit operates at rated load. As a rule, the whistling sound occurs only at medium currents, and with a heavy load, the inductance of the inductor due to the magnetization of the core drops and the whistling stops.

If the whistling sound stops at low currents and with a further increase in the load current, the output transistor begins to heat up sharply, then the area of ​​the magnetic core is insufficient to operate at the selected generation frequency - it is necessary to increase the operating frequency of the microcircuit by selecting resistor R4 or capacitor C3 or install a larger inductor. In the absence of a power transistor of the p-n-p structure, powerful transistors can be used in the circuit n-p-n structures, as it shown on the picture.

As a diode VD5 in front of inductor L1, it is advisable to use any available diodes with a Schottky barrier, rated for a current of at least 10A and a voltage of 50V; in extreme cases, you can use mid-frequency diodes KD213, KD2997 or similar imported ones. For the rectifier, you can use any powerful diodes with a current of 10A or a diode bridge, for example KBPC3506, MP3508 or the like. It is advisable to adjust the shunt resistance in the circuit to the required value. The range of adjustment of the output current depends on the ratio of the resistances of the resistors in the output circuit 15 of the microcircuit. In the lower position of the current control variable resistor slider in the diagram, the voltage at pin 15 of the microcircuit must match the voltage on the shunt when the maximum current flows through it. The variable current control resistor R3 can be set with any nominal resistance, but you will need to select a fixed resistor R2 adjacent to it to obtain the required voltage at pin 15 of the microcircuit.
The variable output voltage adjustment resistor R9 can also have a wide range of nominal resistance 2 ... 100 kOhm. By selecting the resistance of resistor R10, the upper limit of the output voltage is set. The lower limit is determined by the ratio of the resistances of resistors R6 and R7, but it is undesirable to set it less than 1 V.

The microcircuit is installed on a small printed circuit board 45 x 40 mm, the remaining elements of the circuit are installed on the base of the device and the radiator.

The wiring diagram for connecting the printed circuit board is shown in the figure below.

The circuit used a rewound TS180 power transformer, but depending on the magnitude of the required output voltages and current, the power of the transformer can be changed. If an output voltage of 15 V and a current of 6 A is sufficient, then a power transformer with a power of 100 W is sufficient. The radiator area can also be reduced to 100...200 cm2. The device can be used as a laboratory power supply with adjustable output current limitation. If the elements are in good working order, the circuit starts working immediately and only requires adjustment.

Source: http://shemotechnik.ru